The present invention relates to oscillator circuits, and also relates to a magnetron powered by the rectified AC output of an oscillator circuit.
Referring to FIG. 1, which shows a series resonant circuit comprising an inductor L and a capacitor C3, it can be shown that the current in a resistive load LD, which is connected in parallel with capacitor C3, is independent of the load resistance when the current I is oscillating at the resonant frequency and is therefore in phase with the voltage V.sub.X-Y. In such a case the current flowing in load LD can be expressed as V.sub.X-Y /.OMEGA.L where .OMEGA.=2.pi.f where f is the resonant frequency. Thus, at resonance, such a circuit acts as a constant current source and is particularly suitable (after rectification) for powering non-linear loads such as a magnetron which has a zener characteristic.
A resonant converter is disclosed in EP-A-121,917 (published European Patent Application No. 0 121 917) and is used to energize a discharge tube. The principle of operation of this type of resonant converter is illustrated in the accompanying FIGS. 2A to 2F which are circuit diagrams illustrating the oscillating current flows in this type of circuit. FIGS. 2A to 2F have been simplified by omitting the transformers utilized to control switching transistors T.sub.1 and T.sub.2 and by omitting the load which is connected in parallel with capacitor C3.
Referring to FIGS. 2A to 2F, it will be seen that the circuit comprises an inductor L and a capacitor C3 connected in series and gate-controlled switching means, namely, MOSFET's T.sub.1 and T.sub.2, connected to the free terminal of inductor L. The free terminal of capacitor C3 is connected to the positive power supply terminal via a capacitor C1 and to the negative power supply terminal by a capacitor C2. Switching transistor T1 is bypassed by a parasitic diode D1 and switching transistor T2 is bypassed by a parasitic diode D2, both parasitic diodes being poled in the opposite sense to the polarity of the power supply terminals. In each of FIGS. 2A to 2F the flow of current 1 is indicated. Current flow from right to left as shown in FIG. 2A is taken to be flow in the forward direction and, accordingly, current flow as shown in FIG. 2D is flow in the reverse direction.
If switching transistors T.sub.1 to T.sub.2 are switched at the zero crossing points of the current wave form, the circuit oscillates at the resonant frequency as shown in FIG. 4 which is a plot of the current wave form I and voltage wave form V.sub.X-Y (as defined in FIG. 1). It will be seen that the positive half cycles correspond to the current flow shown in FIG. 2A and that the negative half cycles correspond to the current flow shown in FIG. 2D.
In the more general case, as illustrated in FIG. 5, which is a plot of current I and voltage V.sub.X-Y of the circuit of FIGS. 2A to 2F, the switching transistors T.sub.1 and T.sub.2 are not switched at the zero crossing point of the current wave form and accordingly there is a phase difference between the current wave form and the voltage wave form V.sub.X-Y. The periods corresponding to the current flows shown in FIGS. 2A, 2B, 2D and 2E are shown in FIG. 5. Referring now to FIGS. 2A to 2F, when switching transistor T.sub.1 is switched on, current flow in the "forward" direction as shown in FIG. 2A occurs. On switching off transistor T.sub.1, this current flow is continued in a loop through inductor L, capacitor C3, capacitor C2 and parasitic diode D2 as shown in FIG. 2B. The current flow in inductor L and capacitor C3 then reverses. If transistor T.sub.2 is switched on as soon as the zero-crossing point of the current wave form is reached, then the current flow as shown in FIG. 2D occurs. However, in practice there is a slight delay (assumed to be negligible in FIG. 5), so that the initial reversal occurs through a loop consisting of capacitor C1, capacitor C3, inductor L and parasitic diode D1 as shown in FIG. 2C. The current flow shown in FIG. 2D is followed by a current flow in the forward direction through a loop consisting of capacitor C1, capacitor C3, inductor L and parasitic diode D1 as shown in FIG. 2E, as a result of transistor T.sub.2 being switched off. This current flow gradually dies away until the zero-crossing point of the current wave form is reached. Assuming that transistor T.sub.1 is switched on as soon as this zero-crossing point is reached, the forward current flow as shown in FIG. 2A then occurs. However, in practice, transistor T.sub.1 will be switched on shortly after the zero-crossing point and in this intermediate condition between the zero-crossing point and the instant at which transistor T.sub.1 is switched on, current flow in a loop as shown in FIG. 2F occurs, namely through inductor L, capacitor C3, capacitor C2 and parasitic diode D2.
It will be noted that the circuit disclosed in EP-A-121,917 is not controllable and can oscillate only at the resonant frequency as shown in FIG. 4 (already referred to).
However, a paper by Sebastian et al entitled "Regulated Self-Oscillating Resonant Converters", 2nd European Conference on Power Electronics and applications held at Grenoble, France 22-24 Sept. 1987 discloses in FIG. 6 a somewhat complex circuit which automatically switches the switching transistors in such a manner as to achieve non-resonant oscillation as shown in accompanying FIG. 5 (already referred to). By controlling the switching of transistors, the phase difference between the voltage and current wave forms can be controlled and this in turn controls the power output of the circuit.
The circuit disclosed by Sebastian utilizes saturable transformers whose primary windings are connected in series with the inductor and capacitor of the circuit and whose secondary windings are connected across the gate and source of the switching transistors. The drive is removed from the gates of the switching transistors when the saturable transformers saturate. The saturable transformers each incorporate additional windings through which a controlled direct current is fed. This controlled current is generated by an external circuit and generates an additional magnetizing force in each transformer core. It is implicit in the circuits disclosed by Sebastian et al that each transformer core saturates during each half cycle at an instant which is determined by the control current in the additional windings. By varying the control current flowing in the additional windings, the voltage gain and power output of the circuit can be controlled.